This invention is in the field of digital communications, and is more specifically directed to timing recovery from received signals in such communications.
Digital Subscriber Line (DSL) technology has become a primary technology providing high-speed Internet access, and now video and telephone communications, in the United States and around the world. As is well known in the art, DSL communications are carried out over existing telephone “wire” facilities, between individual subscribers and a central office (CO) location, operated by a telephone company or an Internet service provider. Typically, some if not all of the length of the loop between the CO and the customer premises equipment (CPE) consists of conventional twisted-pair copper telephone wire. Remarkably, modern DSL technology is able to carry out extremely high data rate communications, even over reasonably long lengths (e.g., on the order of 15,000 feet) of twisted-pair wire, and without interfering with conventional voiceband telephone communications, carried out over the twisted-pair wire simultaneously with the DSL data communications.
Modern DSL communications achieve these high data rates through the use of multicarrier modulation (MCM) techniques, more specifically discrete multitone modulation (DMT), by way of which the data signals are modulated onto orthogonal tones, or subcarriers, within a relatively wide frequency band (on the order of 1.1 MHz for conventional ADSL, and on the order of 2.2 MHz for ADSL2+), residing above the telephone voice band. The data symbols modulated onto each subchannel are encoded as points in a complex plane, according to a quadrature amplitude modulation (QAM) constellation. The number of bits of data that are carried over each subchannel (i.e., the “bit loading”), and thus the number of points in the QAM constellation for that subchannel, depend on the signal-to-noise ratio (SNR) at the subchannel frequency, which in turn depends on the noise and signal attenuation present at that frequency. For example, relatively noise-free and low attenuation subchannels may communicate data in ten-bit to fifteen-bit symbols, represented by a relatively dense QAM constellation with short distances between constellation points. On the other hand, noisy channels may be limited to only two or three bits per symbol, requiring a greater distance between adjacent points in the QAM constellation to resolve the transmitted symbol. The sum of the bit loadings over all of the subchannels in the transmission band for a DSL link of course amounts to the number of transmitted bits per DSL symbol for that link. And the data rate for DSL communications corresponds to the product of the symbol rate with the number of bits per DSL symbol.
FIG. 1 illustrates the data flow in conventional DSL communications, in a single direction (e.g., downstream, from a central office “CO” to customer premises equipment “CPE”). Typically, each DSL modem (i.e., both at the CO and also in the CPE) includes a transceiver (i.e., both a transmitter function and a receiver function), so that data is also communicated in the opposite direction over transmission channel LP according to a similar DMT process. As shown in FIG. 1, the input bitstream that is to be transmitted, typically a serial stream of binary digits in the format as produced by the data source, is applied to constellation encoder 11 in a transmitting modem 10. Constellation encoder 11 fundamentally groups the bits in the input bitstream into multiple-bit symbols that are used to modulate the DMT subchannels, with the number of bits in each symbol determined according to the bit loading assigned to its corresponding subchannel, based on the characteristics of the transmission channel as mentioned above. Encoder 11 may also include other encoding functions, such as Reed-Solomon or other forward error correction coding, trellis coding, turbo or Low Density Parity Check (LDPC) coding, and the like. The symbols generated by constellation encoder 11 correspond to points in the appropriate modulation constellation (e.g., QAM), with each symbol associated with one of the DMT subchannels. Following constellation encoder 11, shaping function 12 derives a clip prevention signal included in the encoded signals to be modulated, to reduce the peak-to-average ratio (PAR) as transmitted as described in commonly assigned U.S. Pat. No. 6,954,505, issued Oct. 11, 2005, and incorporated herein by this reference.
These encoded symbols are applied to inverse Discrete Fourier Transform (IDFT) function 13, which associates each symbol with one subchannel in the transmission frequency band, and generates a corresponding number of time domain symbol samples according to the Fourier transform. As known in the art, cyclic insertion function 14 appends a cyclic prefix or suffix, or both, to the modulated time-domain samples from IDFT function 13, and presents the extended block of serial samples to parallel-to-serial converter 15. Cyclic insertion function 14 may follow rather than precede parallel-to-serial converter 15 in the transmission sequence, in some implementations. In either case, the time-domain serial sequence, as may be upsampled (not shown) as appropriate, is applied to digital filter function 16, which processes the datastream in the conventional manner to remove image components and voice band or Integrated Services Digital Network (ISDN) interference. The filtered digital datastream signal is converted into the analog domain by digital-to-analog converter 17. After conventional analog filtering and amplification (not shown), the resulting DMT signal is transmitted over a channel LP, over some length of conventional twisted-pair wires, to a receiving DSL modem 20, which, in general, reverses the processes performed by the transmitting modem to recover the input bitstream as the transmitted communication.
At receiving DSL modem 20, analog-to-digital conversion 22 converts the filtered analog signal into the digital domain, following which conventional digital filtering function 23 is applied to augment the function of pre-conversion receiver analog filters (not shown). A time domain equalizer (TEQ) (not shown) may apply a finite impulse response (FIR) digital filter to effectively shorten the length of the impulse response of the transmission channel LP. Frame alignment function 24 receives the sequence of filtered digital samples, and arranges these samples into frames, by removing the cyclic extension from each block of samples, and by performing serial-to-parallel conversion to apply a block of samples (2N) to Discrete Fourier Transform (DFT) function 27. DFT function 27 recovers the modulating symbols at each of the subchannel frequencies, by reversing the IDFT performed by function 13 in transmission. The output of DFT function 27 is a frequency domain representation of the transmitted symbols multiplied by the frequency-domain response of the effective transmission channel. Frequency-domain equalization (FEQ) function 28 divides out the frequency-domain response of the effective channel, recovering the modulating symbols, each representable as a point in a QAM constellation. Constellation decoder function 29 then resequences the symbols into a serial bitstream, decoding any encoding that was applied in the transmission of the signal and producing an output bitstream that corresponds to the input bitstream upon which the transmission was based. This output bitstream is then forwarded to the client workstation, or to the central office network, as appropriate for the location.
It is well known in the art, and will be apparent to the reader from this description, that the demodulation and recovery of data by receiving modem 20 requires synchronized timing with transmitting modem 10. This synchronization includes clock synchronization (in both frequency and phase), and also frame synchronization (synchronization of groups of bits). Clock synchronization is essential so that the sampling of the received signal by ADC 22 is accurate, and remains accurate over many sample periods. Frame synchronization of course ensures that the grouping of digital samples at demodulation by DFT 27, in receiving modem 20, corresponds to the same grouping of digital values that were modulated by IDFT 13 of transmitting modem 10; frame misalignment results in inter-symbol interference (ISI), as known in the art. However, also as known in the art, there is no extrinsic clock signal communicated from transmitting modem 10 to receiving modem 20. Rather, receiving modem 20 must derive the transmit clock from the received signals themselves. As such, receiving modem 20 includes timing recovery function 25, which receives signals from before and after demodulating DFT 27, and that generates a sample clock applied to ADC 22 (and generates other clock signals controlling other functions within receiving modem 20, such as frame alignment function 24).
FIG. 2 illustrates the construction of conventional timing recovery function 25. According to this conventional construction, a phase-locked loop consisting of phase/frequency detector (PFD) 2, low pass filter 4, and voltage-controlled oscillator (VCO) 5 generates sample clock fs from a timing signal based on the demodulated output signal of DFT 27. Typically, as known in the art, PFD 2 generates a phase error signal φ_e based on observed rotations of a demodulated (i.e., DFT) pilot tone within the received signal. For example, a phase or frequency error in the sampling and demodulating of a received signal will appear as a phase rotation in the complex plane; this phase rotation can be detected for the pilot tone, because the QAM-modulated data is known by receiving modem 20. This phase error signal φ_e is filtered by low-pass filter 4 to produce a control signal that is applied to VCO 5. VCO 5 generates and adjusts the timing of sample clock fs in response to the filtered phase error signal.
In addition, conventional timing recovery function 25 includes frame offset estimate function 3. According to conventional approaches, frame_offset estimate function 3 determines the extent of an offset between the frame boundaries detected by frame alignment function 24 and the true frame boundaries. This offset is detectable by analyzing the demodulated pilot tone, or by identifying the cyclic extension in the received signal prior to demodulation, and generates feedback signal frame_offset based on that determination. Frame alignment function 24 adjusts its operation in response to feedback signal frame_offset, to reduce or eliminate ISI. Typically, frame_offset estimate function 3 is operable only during training of modems 10, 20 for the session, based on the assumption that, in showtime, phase and frequency error will not become so large as to lose frame synchronization.
While this conventional timing recovery approach has proven useful in DSL communications, several limitations in its operation have been observed, in connection with this invention. Because only phase error (i.e., phase error signal φ_e) is fed back to VCO 5, the actual phase error may exceed the “pull-in” range of the phase-locked loop of PFD 2, filter 4, and VCO 5. Of course, this phase-locked loop can be constructed to increase its pull-in or lock-in range, but these higher requirements necessarily require more complexity in these circuit functions, and thus substantially higher cost. Furthermore, because only the single pilot tone signal is analyzed by PFD 2 to determine the phase error, corruption of the pilot signal due to noise or attenuation directly affects the precision of the timing recovery. Indeed, if sufficient noise or attenuation is present at the pilot tone frequency, timing may be frequently destroyed, in turn requiring frequent re-initialization of the DSL session.